Level conversion circuit and semiconductor integrated circuit device employing the level conversion circuit

ABSTRACT

In a level conversion circuit mounted in an integrated circuit device using a plurality of high- and low-voltage power supplies, the input to the differential inputs are provided. In a level-down circuit, MOS transistors that are not supplied with 3.3 V between the gate and drain and between the gate and source use a thin oxide layer. In a level-up circuit, a logic operation function is provided.

This application is a continuation application of U.S. Ser. No.13/095,480, filed Apr. 27, 2011, which is a continuation application ofU.S. application Ser. No. 12/169,408, filed on Jul. 8, 2008 (now U.S.Pat. No. 7,944,656), which is a continuation application of U.S.application Ser. No. 11/484,690, filed on Jul. 12, 2006, now U.S. Pat.No. 7,403,361, which is a continuation of U.S. application Ser. No.11/041,232, filed on Jan. 25, 2005, now U.S. Pat. No. 7,091,767, whichis a continuation application of U.S. application Ser. No. 10/647,280,filed Aug. 26, 2003, now U.S. Pat. No. 6,853,217, which is acontinuation application of U.S. application Ser. No. 10/303,841, filedNov. 26, 2002, now U.S. Pat. No. 6,677,780; which is a continuationapplication of U.S. application Ser. No. 10/122,178, filed Apr. 16,2002, now U.S. Pat. No. 6,504,400, which is a continuation applicationof U.S. application Ser. No. 09/833,627, filed Apr. 13, 2001, now U.S.Pat. No. 6,392,439, which is a continuation application of U.S.application Ser. No. 09/209,755, filed Dec. 11, 1998, now U.S. Pat. No.6,249,145, the entire disclosures of all of the above-identifiedapplications are hereby incorporated by reference. This applicationclaims priority to JP 09-359273, filed Dec. 26, 1997.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to semiconductor integratedcircuit devices and level conversion circuits, and more particularly, tosemiconductor integrated circuit devices in which a plurality of circuitunits driven by a plurality of different power supply voltages areformed on a single substrate, and to level conversion circuits used inthe semiconductor integrated circuit devices.

2. Description of the Related Art

The trend in manufacturing semiconductor integrated circuit devices(such as large-scale integrated circuit devices) is to use lower powersupply voltages to reduce power consumption. Recent integrated circuitdevices are driven by 1.2 V power supplies, even though input/outputunits (I/O units), the interfaces with circuits driven by an external3.3 V power supply, are also driven by a 3.3 V power supply.

Additionally, a single semiconductor chip may have two or more circuitblocks that are driven by different respective supply voltages. Suchcircuit blocks require level conversion circuits for raising or loweringvoltage levels between circuit blocks having different respective supplyvoltages. FIG. 1( a) schematically illustrates a circuit diagram of aconventional level-down circuit (a circuit for converting alarge-amplitude signal output by a circuit block operating on a 3.3 Vpower supply, for example, into a small-amplitude signal for input to acircuit block operating on a 1.2 V power supply, for example), and FIG.2( a) schematically illustrates a conventional level-up circuit (acircuit for converting a small-amplitude signal output by a circuitblock operating on a 1.2 V power supply, for example, into alarge-amplitude signal for input to a circuit block operating on a 3.3 Vpower supply, for example).

In FIG. 1( a), VDDQ represents a 3.3 V input, VDD is a 1.2 V powersupply, and VSS a reference, or ground, potential. Thus, VDDQ is alarge-amplitude signal, and the output is a small-amplitude signal basedupon the VDD potential.

In FIG. 1( a), a P-type MOS (PMOS) transistor 200 and an N-type MOS(NMOS) transistor 201 are shown, connected to receive on theirrespective gates an input IN0 having an amplitude of 0.0 V when low and3.3 V when high, for example. IN0 is thus considered to be alarge-amplitude signal input. The circuit shown in FIG. 1( a) outputs asmall-amplitude signal out0 having an output value of 1.2 V, forexample, based upon the power supply VDD. FIG. 1( b) illustrates therespective waveforms of IN0 and out0.

Since, in the PMOS transistor 200 and NMOS transistor 201, a maximumvoltage of 3.3 V may be applied between gate and source, PMOS transistor200 and NMOS transistor 201 are formed with a thick gate oxide layer.

In FIG. 2( a), the level-up circuit is constituted by PMOS transistors202, 203 and NMOS transistors 204, 205. Small-amplitude input signalsin0 and in0 b are complementary dual rail signals. Output signal OUT0 isa large-amplitude output signal of, for example, 3.3 V, based upon powersupply VDDQ. MOS transistors 202-205 each have a thick gate oxide layersimilar to that of the MOS transistors 200, 201 of FIG. 1( a). FIG. 2(b) illustrates the respective waveforms of input signals in0, in0 b andoutput signal OUT0.

In a conventional level-down circuit such as that shown in FIG. 1( a),the logic threshold is typically VDD/2, or close to 0.6 V.Large-amplitude input signals, because their amplitudes are relativelylarge, generally tend to produce noise of a type such that the groundlevel fluctuates. When the ground level fluctuates more than 0.6 V, thesignal is judged erroneously to be a high level in the circuit of FIG.1( a), resulting in a low-level output at out0. Hence, in theconventional level-down circuit, as the VDD supply decreases in voltage,the logic threshold becomes lower, and an incorrect logic value may beproduced at the output out0 in the presence of even very small noise.

In the level-up circuit of FIG. 2( a), when the VDDQ power supply is onbut the input power VDD is off, the values of in0 and in0 b areundefined, causing a through-current to flow between VDDQ and VSS.Hence, in a system where VDD is produced from VDDQ by a DC-DC converter,a heavy load is exerted on the VDDQ power supply, causing a phenomenon,in which the VDD power supply cannot be turned on. If the VDD powersupply cannot be turned on, in0 and in0 b remain undefined, leaving thesystem permanently unable to start normally.

Not only when the power is turned on, but while the VDDQ power supply ison, it is impossible to cut off the VDD power supply because the cutoffof the VDD power renders the values of in0 and in0 b undefined, causinga through-current to flow through the VDDQ and resulting in asignificant increase in power consumption by the system.

Furthermore, the conventional input/output circuit unit that includes anoutput buffer circuit unit also has a similar problem to that discussedabove with respect to the level conversion circuit unit. When the VDDQpower supply is turned on but the VDD power is not, the input signalvalue of the output buffer of the input/output circuit becomesundefined, causing a through-current to flow between VDDQ and VSS of theoutput buffer circuit.

SUMMARY OF THE INVENTION

An object of this invention is to provide a level-down circuit that doesnot readily produce an erroneous output in the presence of ground levelfluctuation in large-amplitude input signals, and to provide asemiconductor integrated circuit device employing the level-downcircuit.

Another object of this invention is to provide a level conversioncircuit in which no through-current flows between a high-voltage powersupply and a ground power supply, and to provide a semiconductorintegrated circuit device employing the level conversion circuit, evenwhen the high-voltage power supply is turned on but the low-voltagepower is not.

Another object of the present invention is to provide a semiconductorintegrated circuit device including a plurality of circuit blockspowered by different respective supply voltage levels, and levelconversion circuits according to the invention for translating voltagelevels between the various circuit blocks.

To achieve these and other objects of the invention, and to solveproblems of the prior art, the present invention includes one or more ofthe following features in the various embodiments discussed in greaterdetail below:

(1) The input to a level-down circuit is provided differentially;

(2) In the level-down circuit, MOS transistors that do not receive 3.3 Vbetween gate and drain or between gate and source have thin gate oxidelayers;

(3) A level-up circuit has a logical operation function; and

(4) An output buffer circuit provided with a level-up circuit includesmeans preventing a through-current from flowing through the outputbuffer when only one of the MOS transistors of the output buffer isturned on.

BRIEF DESCRIPTION OF THE DRAWINGS

FIGS. 1( a) and 1(b) respectively show a circuit diagram of aconventional level-down circuit and its operation waveform diagram.

FIGS. 2( a) and 2(b) respectively show a circuit diagram of aconventional level-up circuit and its operating waveform diagram.

FIGS. 3( a) and 3(b) respectively show a level-down circuit according toa preferred embodiment of the present invention and its operatingwaveform diagram.

FIGS. 4( a) and 4(b) respectively show a circuit diagram of a preferredembodiment of a level-up circuit of the present invention and itsoperating waveform diagram.

FIGS. 5( a) and 5(b) respectively show a circuit diagram of anotherembodiment of a level-up circuit of the present invention and itsoperation waveform diagram.

FIG. 6 is a circuit diagram of a further embodiment of a level-upcircuit of the present invention.

FIGS. 7( a) and 7(b) respectively show a circuit diagram of a furtherembodiment of a level-up circuit of the present invention, and itsoperating waveform.

FIG. 8 is a diagram showing a circuit configured by adding a logicoperation function to the level-up circuit of FIG. 4( a).

FIG. 9 shows an example of providing the level conversion circuit ofFIG. 8 with an output fixing function.

FIG. 10 shows another example of applying the level conversion circuitof FIG. 8 with an output fixing function.

FIG. 11 shows a further example of a level-up circuit having an outputfixing function.

FIG. 12 shows still another example of a level-up circuit having anoutput fixing function.

FIG. 13 shows an example of a level-up circuit having an output fixingfunction of a type that holds the level-converted output.

FIG. 14 shows a system using a level conversion circuit according to thepresent invention.

FIG. 15 shows a system using a level conversion circuit of thisinvention when a circuit block comprising low-threshold MOS transistorsis divided into two.

FIG. 16 shows the system of FIG. 15 with a substrate bias control added.

FIG. 17( a) shows an embodiment for controlling a power switch of FIGS.15 and 16, and FIG. 17( b) shows an example of a method of controllingthe power switch of FIGS. 15 and 16 when a low-threshold MOS transistoris used for the power switch.

FIG. 18 shows an embodiment for generating a gate voltage for theembodiment shown in FIG. 17( a).

FIG. 19 shows an example of an input/output circuit connected to theexternal terminal (pin) of an IC (semiconductor integrated circuit)according to a preferred embodiment of the present invention.

FIG. 20( a) shows an example of an INV used in the embodiment of FIG.19, FIG. 20( b) shows an example of a NAND circuit used in theembodiment of FIG. 19, FIG. 20( c) shows an example of a NOR circuitused in the embodiment of FIG. 19, FIG. 20( d) shows an example of anelectrostatic protective device used in the embodiment of FIG. 19, andFIG. 20( e) shows an example of another electrostatic protective deviceused in the embodiment of FIG. 19.

FIG. 21 shows an example of an input/output circuit that rendersunnecessary circuit portions of FIG. 19 that are substantiallyinoperable.

FIGS. 22( a) and 22(b) respectively show a further embodiment of thecircuit for preventing a through-current from flowing through the outputbuffers PB1 and NB1 at the time of power supply turn-on, and anoperation waveform therefor.

FIG. 23 shows an example of the layout of the input/output circuit ofFIG. 19.

FIG. 24 shows an example of the configuration of an inter-power supplyprotective device.

FIG. 25 shows another example of the configuration of an inter-powersupply protective device.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

In the following description, insulated gate field-effect transistors(FETs) and metal-insulator semiconductor FETs represented by the metaloxide semiconductor FET (MOSFET) are referred to simply as MOStransistors. An N-channel MOS transistor whose majority carriers areelectrons is referred to as an NMOS transistor, and a P-channel MOStransistor whose majority carriers are holes is referred to as a PMOStransistor.

A “threshold voltage” (Vth) qualitatively denotes the voltage differencebetween the gate and the source when the drain current starts to flow.Quantitatively, a measured threshold voltage can be obtained by plottingseveral points in a MOS transistor saturated region in which the draincurrent is expressed by the square curve of the difference between thegate-source voltage and the threshold voltage. The threshold voltagedepends on certain parameters, such as the concentration in thesemiconductor substrate surface where an inversion channel is inducedand the thickness of the gate insulating layer. Where comparisons ofmagnitudes of threshold voltage values are made in the followingembodiments, it should be understood that both PMOS transistors and NMOStransistors operate in enhancement mode, and their threshold voltagevalues are compared as absolute values. If process parameters thatdetermine the channel conductance β are the same, a MOS transistorhaving a greater drain current for the same gate-source voltage may beconsidered to have a lower threshold voltage, assuming that the channelwidth W and the channel length L are the same.

Although the source and drain of a MOS transistor are determinedessentially by the bias of the circuit, in the accompanying drawings,the source of a PMOS transistor is labeled by an arrow pointing towardthe gate electrode, and that of an NMOS transistor with an arrowpointing away from the gate electrode. An electrode whose bias directionchanges during operation (such as a transmission gate) is labeled by abi-directional arrow. When the source and drain are generally notedwithout any distinction, they are called source-drains.

In many integrated circuits, the gates and source-drains of MOStransistors that need large conductances are often commonly connected(the current paths between the sources and drains are connected inparallel) or are distributed equivalently in many cases. In thisspecification, such MOS transistors are represented by a single MOStransistor unless otherwise specifically stated. Likewise, where aplurality of MOS transistors have current paths between source and drainconnected in series and gates applied with the same signal, such MOStransistors are represented by a single MOS transistor in thisspecification unless otherwise stated.

FIG. 3( a) shows a circuit diagram of a level-down circuit according toa preferred embodiment of the present invention. FIG. 3( b) illustratesbasic operation waveforms of the circuit. In FIG. 3( a), 3.3 V(large-amplitude) complementary dual rail input signals are representedby IN0 and IN0B. The 1.2 V (small-amplitude) output signal is denoted byout0. Throughout the specification, and particularly with reference toFIGS. 1-13, signals denoted by capital letters (IN, OUT) are 3.3 V(large-amplitude) signals, and signals denoted by lower-case letters(in, out) are 1.2 V (small-amplitude) signals.

In FIG. 3( a) NMOS transistors 102, 103 have a thick gate oxide layersimilar to that of NMOS transistor 201 shown in FIG. 1( a). PMOStransistors 100, 101 have thin oxide layers by comparison. Voltagesapplied between the gate and drain and between the gate and source ofPMOS transistors 100, 101 are small-amplitude voltages VDD (1.2 V) atmost, and thus the PMOS transistors 100, 101 do not require gate oxidelayers having the large dielectric strength of the gate oxide layers ofNMOS transistors 102, 103, which receive large-amplitude signals. Hence,the PMOS transistors 100, 101 have the smaller gate oxide layerthicknesses, and (though not limited) lower threshold values than thoseof NMOS transistors 102, 103. Using PMOS transistors 100, 101 with thingate oxide layers makes the circuit capable of higher-speed operation.

In this embodiment, because the circuit receives differential inputs atIN0 and IN0B, erroneous logic levels are not output from out0 even inthe presence of ground level fluctuating noise. Moreover, this circuitis not easily influenced by noise even when VDD is lowered.

Another advantage of the present embodiment is that the manufacturingprocess can be simplified by setting the gate oxide layer thickness andthreshold voltage of PMOS transistors 100, 101 equal to those of MOStransistors that form the circuit to which the output out0 is connected,and by setting the gate oxide layer thickness and threshold voltage ofNMOS transistors 102, 103 equal to those of the MOS transistors forminga circuit that provides the inputs IN0, IN0B. For example, NMOStransistors 102, 103 may be output stage MOS transistors of an I/Ocircuit or the MOS transistors used in the protective circuit.

FIG. 4( a) shows an example of a circuit diagram for a level-up circuit,and FIG. 4( b) shows example operation waveforms for the circuit of FIG.4( a). Signals in0 and in0 b represent complementary dual railsmall-amplitude input signals of VDD (1.2 V). The circuit provides a 3.3V (large-amplitude) output at OUT0.

PMOS transistors 300, 301, 302, 303 have thick gate oxide layers similarto PMOS transistor 200 of FIG. 1( a). NMOS transistors 304, 305 alsohave thick gate oxide layers like that of NMOS transistor 201 of FIG. 1(a). As shown in FIG. 4( b), the logic level of in0 is increased inamplitude for output at OUT0. Because of the differential inputs, thiscircuit features strong immunity to noise.

FIGS. 5( a) and 5(b), like FIGS. 4( a) and 4(b), show a level-up circuitdiagram and its associated operation waveforms. However, while thecircuit of FIG. 4( a) converts a 1.2 V-amplitude signal spanning VDD(1.2 V) to VSS (0 V) into a 3.3 V-amplitude signal spanning VDDQ (3.3 V)to VSS (0 V), the circuit of FIG. 5( a) converts a 1.2 V-amplitudesignal spanning VDD (1.2 V) to VSS (0 V) into a 3.3 V-amplitude signalspanning VDD (1.2 V) to VSSQ (−2.1 V). VSSQ is a negative power supplyof −2.1 V. Input signals in0 and in0 b are small-amplitude complementarydual rail input signals. Output OUT0 has a 3.3 V amplitude(large-amplitude) ranging between 1.2 V and −2.1 V. PMOS transistors400, 401, 402, and 403 are thick gate oxide layer transistors similar toPMOS 200 of FIG. 1( a). NMOS transistors 404, 405 are thick gate oxidelayer transistors similar to NMOS transistor 201 of FIG. 1( a).

As shown in FIG. 5( b), the logic level of in0 is increased in amplitudeand output to OUT0. Because of the differential inputs, this circuitfeatures strong immunity to noise, like that of FIG. 4( a).

Since the circuits of FIGS. 4( a) and 5(a) have a complementaryrelationship, features of the level conversion for both embodiments willbe described on the basis of FIG. 4( a) alone. However, such features,including the expansion of the voltage range, are equally applicable tothe circuit of FIG. 5( a), albeit in the negative direction in thecircuit of FIG. 5( a).

FIG. 6 illustrates a level-up circuit which is a modification of thecircuit of FIG. 4( a), to be used at a lower VDD voltage.

FIG. 6 uses an additional PMOS transistor 306 as a current source. Whenthe voltage of VDD is decreased with VDDQ fixed, the “on” currents (thecurrent existing when the potential differences between the source andgate of NMOS transistors 304, 305 are VDD) are smaller than the “off”currents (the current existing when the potential differences betweenthe source and gate of PMOS transistors 302, 303 are VDD). As a result,the cross-coupled PMOS transistors 300, 301 do not provide inversion. Toprevent this, the gate widths of PMOS transistors 300, 301, 302, and 303must be reduced, and the gate widths of NMOS transistors 304, 305increased. Doing so, however, leads to an increased area and increasedinput capacitances respecting the input signals in0 and in0 b. Thus, inFIG. 6, PMOS transistor 306 is connected to the power supply VDDQ. Thisarrangement eliminates the need to reduce the gate widths of PMOStransistors 300, 301, 302, and 303 and to increase the gate widths ofNMOS transistors 304, 305. Only PMOS transistor 306 contributes to anarea increase, keeping the input capacitances respecting the inputsignals from increasing.

Although transistor 306 is shown as a PMOS transistor, it may be an NMOStransistor or any other element for limiting the current. Further, thePMOS transistor 306 may be inserted between PMOS transistors 300 and 302or between PMOS transistors 301 and 303.

FIG. 7( a) shows another modification of the circuit of FIG. 4( a), inwhich an inverter circuit 331 is connected to the output stage of thelevel conversion circuit. Since the output OUT0 of the FIG. 4( a)circuit also serves as an inner node (designated by reference numeral333 in FIG. 7( a)) of the level conversion circuit, the behavior of thevoltage on this inner node may change depending upon the circuitconnected to the output. This affects the delay time of the levelconversion cell, which in turn may cause an erroneous operation. Byinserting the inverter 331 at the output stage as shown in FIG. 7( a),the circuit connected to the output of the level conversion circuit isprevented from adversely affecting the node in the level conversioncell. Further, because the output impedance at OUT0 can be reduced,compared with that of FIG. 4( a), the total delay time when a largenumber of circuits are connected to OUT0 can be reduced.

When the level conversion cell is to be registered by an automaticarranging/routing tool, by use of the configuration of FIG. 7( a) ahigh-speed level conversion cell having excellent noise resistance canbe configured. Further, because the dependence of delay on the load ofthe output is the same as that of the CMOS inverter, the dependence ofthe CMOS can be directly applied to the timing analysis.

FIG. 7( b) is a waveform diagram for the circuit of FIG. 7( a).Inserting the inverter 331 increases the through-rate of the outputOUT0, whereas the through-rate of the inner node 333 itself is slow.

Adding the inverter circuit to the output of the circuit of FIG. 3( a)can also produce a similar effect. Moreover, in the embodimentsdiscussed below, the inverter can be added to the output circuit,although its addition is not specifically mentioned.

FIG. 8 shows a circuit configured by adding a logic operation functionto the level-up circuit of FIG. 4( a). Signals in0 and in1 are 1.2 V(small-amplitude) input signals, and in0 b and in1 b are theircomplementary signals. The circuit outputs a 3.3 V (large-amplitude)output signal OUT0. Compared with FIG. 4( a), the inverter comprisingMOS transistors 302 and 304 and the inverter comprising the MOStransistors 303 and 305 are replaced by a NOR circuit comprising MOStransistors 502, 504, 506, and 508, and by a NAND circuit comprising MOStransistors 503, 505, 507, and 509. With this arrangement provides thelogic operation OUT0=in0 OR in1.

If the NOR circuit comprising MOS transistors 502, 504, 506, and 508 isreplaced with a logic circuit that performs an operation LOG1 and acircuit complementary to the LOG1 circuit is replaced with the NANDcircuit comprising MOS transistors 503, 505, 507, and 509, a level-upcircuit having the logic operation function OUT0−LOG1 (where “−”represents an inversion) results. Further, while the circuit illustratedin FIG. 8 has two inputs (four inputs when the complementary signals areconsidered), a circuit configuration having a greater number of inputsmay be constructed.

FIG. 9 illustrates a circuit configured by providing the level-upcircuit of FIG. 8 with an output fixing function. The level-up circuitwith the output fixing function is designated by reference numeral 513.Furthermore, an inverter 512 is provided as shown, input signal in1 b isreplaced with a 3.3 V (large-amplitude) signal IN1, and in1 is derivedfrom signal IN1 by using the inverter 512.

In FIG. 9, a circuit block 510 operates on a power supply voltage of 1.2V, and a circuit block 511 operates on a power supply voltage of 3.3 V.Thus, the level-up circuit 513 functions to translate from the circuitblock 510 to the circuit block 511. Setting IN1=0 V results in OUT0=3.3V regardless of the voltage signals in0 and in0 b. In this state, nothrough-current flows from the power supply VDDQ to VSS of the level-upcircuit 513.

The power supply of the circuit block 510 can be turned off by settingIN1 =0 V. At this time, although the input signals in0 and in0 b areundefined, no through-current flows through the level-up circuit 513,and its output OUT0 is determined, so that the circuit block 511 doesnot operate erroneously.

When the circuit block 510 is constructed of low-threshold MOStransistors, a subthreshold leakage current flows, consuming power evenduring standby, when the circuit block is not operated. By adopting theconfiguration of FIG. 9, however, the power supply of the circuit block510 can be off during standby, thus suppressing the power consumptiondue to the subthreshold leakage current.

FIG. 9 does not expressly show such circuit constants as the gate widthsof the MOS transistors. Since a large-amplitude signal is input at IN1,the gate lengths of MOS transistors 503, 509, 504 and 508 should be setsmaller than the gate lengths of MOS transistors 505, 507, 502, and 506.Furthermore, although the level conversion circuits discussed below alsodo not expressly show the circuit constants, if the CMOS circuit isconstructed of MOS transistors having large-amplitude inputs (such asMOS transistors 503, 509, 504, and 508) and MOS transistors havingsmall-amplitude inputs (such as MOS transistors 505, 507, 502, and 506),the symmetry of circuit configuration can be maintained by setting thegate lengths of the MOS transistors supplied with the large-amplitudeinputs smaller than the gate lengths of the MOS transistors suppliedwith the small-amplitude inputs.

The level-up circuit 514 shown in FIG. 10 has an output fixing functionfor fixing its output to OUT0=0 V when IN1=3.3 V, by locating theinverter 512 as shown. Furthermore, the output of the FIG. 10 circuit istaken from the node common to the drains of MOS transistors 506, 508,and 504, to which the gate of MOS transistor 501 is also connected.Otherwise, the circuit configuration shown in level 10 is substantiallysimilar to that of FIG. 9. Therefore, when it is necessary to fix theoutput at OUT0=3.3 V, the level-up circuit 513 of FIG. 9 is used; andwhen it is necessary to fix the output at OUT0=0 V, the level-up circuit514 of FIG. 10 is used.

FIGS. 11 and 12 respectively illustrate circuits that realize thefunctions of FIGS. 9 and 10 with a different construction. Level-upcircuit 515 and 516 of FIGS. 11 and 12, respectively, have an outputfixing function. When the power supply of the circuit block 510 isturned off, no through-current flows between the power supplies ofcircuits 515, 516 when input IN1 is set to an appropriate level, therebystabilizing the output OUT0.

In each of FIGS. 9-12, level-up circuits having an output fixingfunction have been shown, whereby the output OUT0 is fixed to apredetermined level. Combining each of these circuits with a latchcircuit forms a circuit that holds the output level OUT0 when IN1becomes a predetermined value.

FIG. 13 shows a preferred example. Level-up circuit 513 of FIG. 9 isshown, with a latch circuit 522 at its output. When IN1 changes from 3.3V to 0 V, the latch circuit 522 latches the signal level of the output521 of level-up circuit 513 to OUT0. When IN1 is 0 V as described above,the power supply of the circuit block 510 can be turned off. Although atthis time the voltages of inputs in0 and in0 b become undefined, nothrough-current flows in the level-up circuit 513 and its output OUT0 isdetermined, so that the circuit block 511 does not operate erroneously.

The latch circuit 522 can also be applied to the level-up circuit shownin FIGS. 10-12 in a similar manner, providing a similar effect.

FIG. 14 shows an example of a circuit system employing theabove-described level-up circuits having an output fixing function, andlevel-down circuits. A low-voltage circuit block 601 is supplied withVDD=1.2 V, and constructed of low-threshold MOS transistors. Ahigh-voltage circuit block 602 is supplied with VDDQ=3.3 V, andconstructed of MOS transistors having a higher threshold than that ofthe MOS transistors forming circuit block 601. Hence, the subthresholdleakage current flowing between power supplies in the circuit block 602is negligible compared with that of the circuit block 601. Level-upcircuits 6031 to 603 n (such as those shown in FIGS. 9-14) having anoutput fixing function and level-down circuit 6041 to 604 n (such asthat shown in FIG. 3( a)) are used to transfer signals between thecircuit blocks 601, 602.

Because the circuit block 601 is constructed with low-threshold MOStransistors, a subthreshold leakage flows, consuming power even duringstandby mode when the circuit block 601 is not being operated. Byinputting an appropriate value by each IN1 of a group of level-upcircuits 603 during standby, however, the power supply of the circuitblock 601 can be turned off, suppressing the power consumption due tothe subthreshold leakage current. Further, because the outputs OUT0 ofthe level-up circuits 603 are fixed, the circuit block 602 does notoperate erroneously.

Although the circuit functions incorporated in the circuit block 602 arenot limited, the circuit block 602 may include circuits having a clockfunction and memories whose power supplies cannot be turned off, therebyallowing the power supply of the circuit block 601 to be turned offfrequently. For turning off the power supply of the circuit block 601, aPMOS, for example, may be inserted between the circuit block 601 and thepower supply VDD. Integrating the circuit system 600 in a single chipeliminates the need to provide a switch outside the chip for turning offthe circuit block 601.

FIG. 15 shows a preferred embodiment wherein the circuit block 601 isdivided into two systems, circuit block 601 a and circuit block 601 b.

The circuit block 601 as shown in FIG. 14 has a drawback that when itspower supply is turned off, the voltages on the nodes inside the circuitblock 601 become undefined, and information contained in memory circuits(such as SRAMs and DRAMs, if any) in the circuit block 601 cannot bemaintained.

In FIG. 15, circuits, such as memories, whose power supplies cannot beturned off are incorporated in the circuit block 601 a, while circuitswhose power supplies may be turned off are incorporated in the circuitblock 601 b. A power switch control circuit PSC is provided to turn onor off power switch PMOS transistors 702 a, 702 b with signals 701 a,701 b from the PSC. Level-up circuits 603 a and 603 b have an outputfixing function, and level-down circuits 604 a and 604 b are alsoprovided. A fixing circuit is preferably inserted between circuit blocks601 a and 601 b to prevent the erroneous operation of the circuit block601 a when the power supply for the circuit block 601 b is turned off;however, the fixing circuit is not shown. It can be easily realized byusing CMOS circuits such as a NAND and NOR.

The system configuration of FIG. 15 has two standby states. One is astate in which the power switch PMOS transistor 702 b is turned off toturn off the power supply of the circuit block 601 b (standby 1). Theother is a state in which, in addition to standby 1, the power switchPMOS transistor 702 a is also turned off to turn off the power supply ofthe circuit block 601 a (standby 2). Standby 1 can reduce thesubthreshold leakage current of the circuit block 601 b. The circuitblock 601 b, because it does not incorporate such circuits as memories,is free from erroneous operation when its power supply changes from “on”to “off”. Hence, recovery from standby 1 can be accomplished at highspeed. On the other hand, when the standby state shifts to standby 2where the power supply of the circuit block 601 a is off, the contentsof the memories in the circuit block 601 a are erased, and consequentlythe recovery from standby 2 takes time. However, standby 2 can reducethe subthreshold leakage current of the circuit block 601 a in additionto setting up standby 1, thus achieving lower power consumption. If theoperation of the circuit blocks 601 a and 601 b is stopped for arelatively short period of time, the standby state should be standby 1.When the operation is stopped for a long period, the standby stateshould be standby 2.

FIG. 16 shows an embodiment in which substrate bias control circuitsVBCa and VBCb are added to the circuit of FIG. 15. As described above, asubthreshold leakage current flows in the circuit block 601 a duringstandby 1. The substrate bias control circuit VBCa controls thesubstrate voltage of the MOS transistors in the circuit block 601 aduring standby 1 as follows:

(1) For PMOS transistors, the substrate voltage is controlled at a levelhigher than the power supply voltage.

(2) For NMOS transistors, the substrate voltage is controlled at a levellower than the power supply voltage.

This control raises the threshold voltage of the MOS transistors in thecircuit block 601 a, and reduces the subthreshold leakage current.Because the power supply remains turned on, the contents of the memoriesin the circuit block 601 a are maintained.

The substrate bias control circuit VBCb connected to the circuit block601 b can be used during an IDDQ test. During the IDDQ test, a circuitto be measured is cut off from a power supply line, and thus the powerswitches PMOS 702 a and 702 b cannot be turned off. The use of thesubstrate bias control circuits VBCa and VBCb, which raise the thresholdvoltage of the MOS transistors forming the circuit blocks 601 a and 601b to reduce the subthreshold leakage current, allows the IDDQ test to beexecuted.

The use of the substrate bias control circuits VBCa, VBCb is not limitedto the circuit configuration of FIG. 16, but can be applied to anysystem which comprises a first circuit block constructed ofhigh-threshold MOS transistors and supplied by a large-amplitudevoltage, and a second circuit block constructed of low-threshold MOStransistors and supplied with a small-amplitude voltage, and in whichthe first and second circuit blocks interface with each other vialevel-up circuits with an output fixing function and level-downcircuits. The first circuit block may incorporate circuits which need tooperate at high-speed, and the second circuit block may include circuitsthat can operate at low speed and do not consume much power, such as anRTC. The first circuit block is divided into circuit blocks 1A and 1B,the circuit block 1A containing circuits such as a memory that takestime for recovery when the power supply is turned off, and the circuitblock 1B containing other circuits. These divided circuit blocks 1A, 1Bcontrol their power supplies and incorporate a substrate bias controlcircuit.

FIG. 17( a) shows an embodiment for controlling the power switch PMOS702 a used in FIGS. 15 and 16. In FIG. 17( a), the power switch 702 a isa high-threshold PMOS transistor. When the transistor is active, thevoltage 701 a on the gate terminal is controlled at a negative value aslong as the dielectric strength of the gate oxide film permits. Thisenables a large current to flow through the PMOS transistor. Thenegative voltage to be applied may be, for example, a negative voltageused for the substrate bias control. In the standby (inactive) state,the gate voltage 701 a is controlled at 1.2 V (VDD). Because the powerswitch PMOS transistor 702 a is a high-threshold MOS transistor, thisgate voltage is high enough to turn off the power switch PMOS 702 a.

FIG. 17( b) shows an embodiment for controlling a low-threshold PMOStransistor power switch 702 a. When active, the gate voltage 701 a ofthe power switch PMOS transistor 702 a is controlled at 0 V. Because thepower switch PMOS transistor 702 a is a low-threshold MOS transistor, alarge current can flow. In the standby state, the gate voltage 701 a iscontrolled at a positive value as long as the dielectric strength of thegate oxide film permits. Here, it is illustratively controlled at 3.3 V,and the power switch PMOS 702 a, although a low-threshold MOStransistor, can have a satisfactory on-off characteristic.

The control shown in FIGS. 17( a) and 17(b) is not limited to PMOStransistor control, but can likewise be applied with an NMOS powerswitch to produce the same effect, except that the polarity is inverted.

FIG. 18 shows an embodiment for generating the gate voltage 701 a shownin FIG. 17( a). A negative voltage generating circuit 710 generates −2.1V from 3.3 V (VDDQ) and outputs it at 712 to a power switch controlcircuit 711. The power switch control circuit 711, which controls thegate voltage 701 a, is also supplied with VDD (1.2 V). The −2.1 V supplyvoltage 712 is also the substrate bias provided to the circuit block 601a for control of the substrate voltage of its MOS transistors, via VBCa.By commonly using the negative supply voltage 712 for the substrate biascontrol and for the control of the power switch 702 a enables asignificant reduction in the size of the circuit required to realize thecontrol of FIG. 17( a).

Next, an example of an input/output circuit using the above-mentionedconversion circuits and connected to an external terminal (pin) of theIC (semiconductor integrated circuit) will be described with referenceto FIG. 19.

In FIG. 19, symbols PB1 and NB1 denote PMOS transistors and NMOStransistors, respectively, both having conductances sufficiently high todrive the load of an external circuit to be connected to externalterminals I/O. Both PB1 and NB1 constitute an output buffer circuit. Aninverter INV7, a NAND gate NAND1, and a NOR gate NOR1 constitute acircuit that performs a tristate logic operation by which, when anoutput control signal /OE is “0”, the information of the output signalOut is led through an output buffer to the external terminal I/O (a MOStransistor in the output buffer is turned on to bring the output bufferto a low output impedance state), and in which, when /OE is “1”, bothMOS transistors of the output buffer are turned off regardless of thestate of the output signal Out to bring the output buffer to a highoutput impedance state.

The external terminal I/O is also connected to the input side of a NORgate NOR2 and used as a common terminal for input and output. When inputcontrol signal /IE is logic “0”, the NOR gate NOR2 transfersinformation, which has been supplied to the external terminal I/O fromthe outside of the IC, to a terminal /In (the /In terminal is theinverted level of a signal supplied to the external terminal I/O), and,when the input control signal /IE is logic “1”, blocks the transfer ofthe information (the /In terminal is forcedly held at logic “0”).

P3 is a pull-up PMOS transistor which is used to supply the externalinput—which takes on either a logic “0” or an open state (high impedancestate)—to the I/O terminal. When pull-up control signal /PU is logic“0”, P3 conducts to transfer to the NOR gate NOR2 a signal of logic “0”when the external input is logic “0”, and a signal of logic “1” when theexternal input is in an open state. The channel length of transistor P3is set larger than its channel width W so that the impedance of P3 whileit conducts is sufficiently larger than that while the external input is“0”.

A low-voltage power supply circuit block is shown at the lefthand sideof FIG. 19 within a dotted-line rectangle, in which, in the range shown,an N-type substrate (N-type well) N-SUB for all PMOS transistors isconnected to a PMOS well power supply Vbp and a P-type substrate (P-typewell) P-SUB for all NMOS transistors is connected to an NMOS well powersupply Vbn. The supply voltages are Vss (0 V) and Vdd (1.2 V). Almostall MOS transistors have lower threshold voltages than that of ahigh-voltage power supply circuit described below, and the gateinsulating layers are thin. The minimum channel length of this circuitblock is, for example, 0.2 μm, which is shorter than 0.32 μm, theminimum channel length of the high-voltage power supply circuit.

For the inverter circuits INV4-INV9, the circuit of FIG. 20( a) may beused, and for the NAND circuit NAND1 and the NOR circuit NOR1, thecircuits of FIGS. 20( b) and 20(c), respectively, may be used.

A high-voltage power supply circuit block is shown at the righthand sideof FIG. 19 enclosed by a dotted-line rectangle. This circuit block haspower supply voltages Vssq (0 V) and Vddq (3.3 V). In the range shown inthe drawing, an N-type substrate (N-type well) N-SUB for all PMOStransistors is connected to the power supply Vddq and a P-type substrate(P-type well) P-SUB for all NMOS transistors is connected to the powersupply Vssq. All MOS transistors have a high threshold voltage and thickgate insulating layers. Although the power supplies Vss and Vssq may beconnected together outside the IC (for example, on the printed circuitboard on which the IC is mounted), their external terminals (pins),bonding pads, and internal circuits inside the IC are separated toprevent variation of the load current from entering the power supplywiring and causing operational noise.

Symbol LSD in the low-voltage power supply circuit denotes a level shift(level-down) circuit that converts a high-amplitude signal of 3.3 V,supplied through the high-voltage power supply circuit, into alow-amplitude signal of 1.2 V that can be processed in the low-voltagepower supply circuit. The LSD may be the circuit shown in FIG. 3( a), inwhich MOS transistors 102 and 103 have thick gate insulating layers thatare preferably formed by the same gate oxide layer forming process usedto form the MOS transistors of the high-voltage power supply circuit.The channel length of the MOS transistors 102 and 103 is the minimumchannel length of the high-voltage power supply circuit (0.32 μm), notthe minimum length of the low-voltage power supply circuit of (0.2 μm).

The MOS transistors of the low-voltage circuit of FIG. 19, in the rangeshown, have thin gate insulating layers except for the level-downcircuit LSD, and have channel lengths equal to the minimum channellength of the low-voltage power supply circuit of (0.2 μm).

LSU1-LSU4 in the high-voltage power supply circuit are level shiftcircuits for raising the level of the 1.2 V low-amplitude signalsupplied from the low-voltage power supply circuit to a high-amplitudesignal of 3.3 V, using the circuit shown in FIG. 4( a) or any of theother level-up circuits described herein, for example.

INV1 and INV2 constitute a pre-buffer circuit to drive the outputbuffers PB1, NB1. INV1 and INV2 may be constituted by the inversioncircuit shown in FIG. 20( a). The output buffers PB1, NB1 are formed ina large area so as to have low output impedances, and hence their input(gate) capacitances are large. The pre-buffers have the following roleand configuration. The pre-buffers reduce the load capacitances of thelevel shift circuits LSU1, LSU2, and the setting of the designparameters of the level shift circuits is not restricted by the largeinput capacitances of the output buffers.

(2) The ON impedances of the PMOS transistors (e.g., PMOS transistors300-303) on the cross-coupled side is set larger than that of the NMOStransistors (e.g., NMOS transistors 304, 305) on the input side, so thatthe previous output states of the level shift circuits LSU1, LSU2 can beinverted by the input signals I and II. To directly drive the outputbuffer by reducing the impedance on the cross-coupled side, theimpedance of the input MOS transistor must be further reduced, which isnot advantageous in terms of the area occupied and the powerconsumption. Hence, the roles are so allocated that the level conversionfunction is performed by the level shift circuit and the output bufferis driven by the pre-buffer. When the input side has NMOS transistors,the output impedance of each circuit when outputting a logic “1” is sodetermined as to be increased, in ascending order, for the outputbuffer, the pre-buffer, and the level shift circuit. The outputimpedance of each circuit when outputting a logic “0” is determined inmost cases in the same order. Considering the switching characteristicsof the output buffer described later, the output impedance may be sodetermined as to be increased, in ascending order, for the outputbuffer, the level shift circuit, and the pre-buffer. Similarly, when theinput side has PMOS transistors, the output impedance of each circuitwhen outputting a logic “0” is so determined as to be increased, inascending order, for the output buffer, the pre-buffer, and the levelshift circuit. Although the output impedance of each circuit whenoutputting a logic “1” is determined in the same order in most cases,the order of impedance may be changed to the ascending order of theoutput buffer, the level shift circuit, and the pre-buffer, consideringthe switching characteristics of the output buffer described later.

(3) When the output buffer shifts from the previous output state to theinverted state, the simultaneous turn-on of both MOS transistors shouldbe avoided, or at least the period during which they both conduct shouldbe short. That is, it is desirable that both MOS transistors be turnedoff relatively early and turned on relatively late. The waveform of thesignal to be fed to the output terminal I/O is preferably made gradualto some degree because too steep a trailing or leading edge of thesignal waveform is likely to induce differential noise in thesurrounding external pins and in the wiring around the printed circuitboard. Considering these points, the output impedances of thepre-buffers are determined.

MOS transistors N1 and P1, whose drains are connected to the input sideof the pre-buffer, prevent a large through-current caused by thesimultaneous turn-on of the buffer MOS transistors PB1 and NB1, whichcan occur because the signal from the low-voltage power supply circuitis not defined when the power supply voltage Vddq is already establishedbut the power supply voltage Vdd is not yet established (the powersupply voltage turn-on sequence is so determined that Vddq isestablished earlier than Vdd), such as may occur when turning on thepower supply for an applied system. P1 conducts when the gate voltage ofPB1 is at a low level “L”, and N1 conducts when the gate voltage of NB1is at a high level “H”. Assuming a normal operation, in the high outputimpedance mode when PB1 and NB1 are both off, N1 and P1 are also bothoff, thus exerting no influence on the normal operation. In the lowoutput impedance mode when only one of PB1 and NB1 is on, the transistorN1 or P1 which is on acts to turn off the other that has been off, thusactually having no effect on the normal operation. In normal operation,PB1 and NB1 cannot both be on, and thus the input voltages of anabnormal state (that is, when the gate voltage of PB1 is low and thegate voltage of NB1 is high) are not supplied. When the signal from thelow-voltage power supply circuit is undefined in the above case, such anabnormal state may occur. However, as the state approaches an abnormalstate, N1 and P1 begin to conduct and act to change the gate voltages ofPB1 and NB1 in the same direction, so that finally only one of PB1 andNB1 is turned on.

MOS transistors N2-N5 provide greater assurance that through-currentwill be prevented during power turn-on in the above case. When the powersupply is turned on and accordingly the outputs Q and /Q of the levelshift circuit LSU1 begin to rise, N3 starts to conduct, pulling theinput /I toward the low level and the output Q toward the high level.Likewise, N2 also begins to conduct, pulling the output /Q toward thelow level and the output Q toward the high level. That is, N2 and N3both act to pull the output Q of the level shift circuit LSU1 toward thehigh level when the power supply is turned on. During the normaloperation, when the input I is high, the output Q is high. At this timeN2 and N3 act to move the output Q to the high level, i.e., in the samedirection. Further, when the input I is low, N2 and N3 are off. Hence,N2 and N3 have no adverse effects on the logic operation of the outputsQ and /Q based on the inputs I and /I.

N4 and N5 operate in a way similar to N2 and N3, and thus theirdescription will be omitted. The only difference is that the connectionto the input and output terminals of the level shift circuit LSU isopposite to that of the level shift circuit LSU1, and hence the output Qis pulled to the low level at the time of power supply turn-on.

Because, at the time of power turn-on, N2-N5 pull the output Q of theLSU2 toward the low level and the output Q of LSU1 toward the highlevel, they both act to turn off the output buffers PB1 and NB1. Hence,if, at the time of power supply turn-on, N1 and P1 operate earlier, onlyone of the output buffers PB1 and NB1 is turned on. If N2-N5 operateearlier, both of the output buffers PB1 and NB1 are turned off. Ineither case, the output buffers PB1 and NB1 can be prevented fromturning on simultaneously.

N6 similarly pulls the output Q of the level shift circuit LSU3 to thehigh level when the power supply is turned on, thereby preventing thestate of the input/output terminal I/O from being transmitted to theinternal circuit /In. Moreover, N7 pulls the output Q of the level shiftcircuit LSU4 to the high level when the power supply is turned on, thusturning off the pull-up transistor P3.

One of N4 and N5 connected to LSU2 and one of N2 and N3 connected toLSU1 may be omitted as in LSU3, LSU4.

ESD1 and ESD2 are electrostatic breakdown protective circuits as shownin FIG. 20( d), for example, which prevent the gate insulating layers ofthe output buffers PB1, NB1 from breaking down when a surge voltageenters the input/output terminal I/O.

Referring back to FIG. 19, a resistor R1 and MOS transistors P2 and N8constitute a circuit for preventing the MOS gate insulating layer of theNOR gate NOR2 from breaking down when a surge voltage enters theinput/output terminal I/O. Resistor R1 and MOS transistor N9 constitutea circuit to prevent the gate insulating layer of pull-up transistor P3from breaking down when a surge voltage enters the input/output terminalI/O.

ESD3-ESD10 are electrostatic breakdown protective circuits, and may beconstructed as shown in FIG. 20( e). These circuits prevent the gateinsulating layers of the level shift circuits LSU1-LSU4 from breakingdown when a surge voltage enters between different power supplies Vddand Vddq, between Vdd and Vssq, between Vss and Vddq, or between Vss andVssq (Vss and Vssq are typically connected on the printed circuit boardwhen the IC is mounted on the board but are open when the IC is handledas a single device, where there is a particular need for measuresagainst surges), and flows through the low-voltage power supply loadcircuit on the left side and the high-voltage power supply load circuiton the right side of FIG. 19. In the circuit of FIG. 20( e), a resistorR3 relaxes the waveform of a surge voltage at I in cooperation with theparasitic capacitor, and also produces a voltage drop when a bypasscurrent flows through a protective device N16 or P16, thereby limitingthe surge voltage impressed on the output terminal O connected to theMOS gates of the level shift circuit LSU1-LSU4. When a surge makes thepotential of the node I is more positive than the power supply Vddq, thesource junction (PN junction) of P16 connected to the node I side isbiased forwardly to form a surge bypass between the node I and the powersupply Vddq through the N substrate (N well) connected to the junctionand the power supply Vddq. When a surge renders the node I more negativethan the power supply Vddq, the drain junction (PN junction) of P16connected to the node I side breaks down in the reverse direction toform a surge bypass between the node I and the power supply Vddq throughthe N substrate (N well) (or further through the source junction on theopposite side) connected to the junction and the power supply Vddq. Thegate of P16 is connected to the power supply Vddq, so that the electricfield concentration is large in the drain junction, lowering theabsolute value of the breakdown voltage.

When a surge voltage is impressed between the node I and the powersupply Vssq, N16 forms a bypass between the node I and the power supplyVssq in a positive-negative relation, contrarily to the case describedabove.

In the normal operation, the above drain junction between P16 and N16 onthe side of the node I is not biased forwardly, nor is it applied with areverse bias over the breakdown voltage. Further, P16 and N16 have theirgates and sources short-circuited and therefore are off. Hence, theprotective circuits do not affect the normal logic operation.

The electrostatic breakdown protective devices described above areprovided in the high-voltage power supply circuit block enclosed of FIG.19. The gate insulation layers are formed thick to prevent theprotective devices themselves from breaking down.

The input/output circuit shown in FIG. 19 is preferably arranged as astandard circuit around bonding pads of multiple chips. According to theuse and kind of the IC, the input/output terminal I/O may be used forinput only or output only, or for both input and output. Unnecessaryinput/output circuits can be made substantially inoperable by theembodiment shown in FIG. 21. C1-C10 denote “broken line” points forrendering a particular circuit of the high-voltage power supplyinoperable by not providing the wiring between the low-voltage powersupply circuit and the high-voltage power supply circuit. S1-S10 showsthat inputs are fixed to a particular logic with low impedance when theinput paths are cut off in such a form. S1-S10 are connected to Vssq(down arrow) or Vddq (up arrow) via the internal wiring of the IC. Whenthe terminal I/O is used, for example, as an input-only terminal, thelines are cut off at points C7-C10 (no wiring pattern is provided) andthe inputs I and /I of the level shift circuits LSU1, LSU2 are connectedto the power supplies as shown to render both the output buffers PB1 andNB1 off. With the inputs of the level shift circuits fixed to aparticular logic level, the buffers do not perform switching, thuspreventing erroneous operation and waste of electric power. By fixingthe inputs of the preceding stage circuits as much as possible, it ispossible to eliminate the need for additional complexity of the circuitsof the succeeding stage.

FIG. 22( a) shows another embodiment of a circuit for preventing athrough-current that may flow through the output buffers PB1 and NB1 atthe time of power supply turn-on. In the figure, parts identical withcorresponding parts of FIG. 19 are designated by like reference symbols.Symbol OG denotes a one-shot pulse generation circuit that generatespulses OSP for a particular period of time after the power supply Vddqis turned on, as illustrated in FIG. 22( b). After the power supplyturn-on, this pulse OSP turns on MOS transistors N1 and P1, bringing theoutputs of the inverters INV1 and INV2 to a low level and a high level,respectively, and turning off both the output buffers PB1 and NB1 at thesucceeding stage. Connecting this one-shot pulse generation circuit OGcommonly to the similar portions of other input/output circuits (throughbuffers) enables compact integration of the input/output circuits andalso makes it possible to set the initial state of the level shiftcircuits LSU1-LSU4 at the time of power supply turn-on.

FIG. 23 shows one preferred embodiment of the layout of the input/outputcircuit shown in FIG. 19.

As shown in FIG. 23, a plurality of I/O pads 2202 are arranged inparallel along a chip end portion 2201. Circuits shown in FIG. 19 arearranged near the chip end side in a direction perpendicular to the chipend side. NMOS buffer 2203 and PMOS buffer 2204 are the MOS transistorsNB1 and PB1 of the output buffers of FIG. 19 and arranged by the side ofthe I/O pads as shown. Arranged toward the inside of the chip are theelectrostatic breakdown protective circuit ESD1 and ESD2 (2205), thepull-up circuit (2206), the pre-buffer (2207), the level shift circuit(2208), and the tristate logic operation circuit (2209).

Power supply wiring is laid on third and fourth metallic wiring layersto extend between the adjoining circuit blocks in a direction parallelto the chip end side. Vssq and Vddq are wired on 2203, Vssq and Vddq on2204, Vssq on 2205, Vddq on 2206, Vssq on 2207, Vddq on 2208, and Vssand Vdd on 2209.

Next, the configuration of an inter-power supply protective device willbe described that can suitably be applied to a chip that, like thesemiconductor integrated circuit device of this invention, uses aplurality of power supply voltages. The semiconductor integrated circuitdevice of this embodiment employs, in particular, a triple wellconstruction. A particularly efficient configuration of the inter-powersupply protective device of the triple well construction will bedescribed in the following.

In chips that use a plurality of power supplies of different voltages(or even power supplies of the same voltage provided separately,depending on the magnitude of power supply noise), there are severalkinds of power supply pins. To allow static electricity to escape easilyand thereby improve the electrostatic dielectric strength in such chips,it is effective to insert such devices as MOS transistors and diodesbetween power supplies and ground and between different power supplies.In this case, connections should be made so that no current flows in theforward direction under a bias present in the normal use condition, butalso so that a current flows in the reverse direction only when staticelectricity of several hundred to several thousand volts enters thechip.

In the case of a triple well structure, a diode can be fabricated infour different ways: between a P-type substrate and an N-type elementregion, between an N-type element region and a P-type well, between aP-type well and an N-type diffusion layer, and between an N-type welland a P-type diffusion layer. The method by which the area is minimizedand the parasitic element effect is small depends on the kind of powersupply to be connected to it.

A particularly efficient configuration of such a protective device ofthe embodiment of this invention will be described below.

FIG. 24( a) shows an example of a particularly efficient way of forminga diode when the diode connections shown in FIG. 24( b) are made in achip having a P-type silicon substrate and supplied with VSS.

FIG. 24( a) shows a silicon substrate (P-type) 2301, an elementformation region (N-type) 2302, an N-type well 2303, a P-type well 2304,an N-type diffusion layer 2305, a P-type diffusion layer 2306, a diode2307 formed by a P-type well formed on the P-type substrate and theN-type diffusion layer 2305, a diode 2308 formed by the N-type well 2303formed on the N-type device formation region 2302 (biased by VDDQ) andthe P-type diffusion layer 2306, a diode 2308 a formed by the N-typewell 2303 formed on the N-type device formation region 2302 (biased byVDD) and the P-type diffusion layer 2306, a diode 2309 formed by theP-type well 2304 formed on the N-type device formation region 2302 andthe N-type diffusion layer 2305, and a diode 2310 formed by an N-typewell formed on the P-type substrate 2301 and the P-type diffusion layer2306.

In the case of a chip where the silicon substrate is of P-type andsupplied with VSS, first, the diode connected to VSS is desirably formeddirectly on the P-type substrate by using the P-type well, the sameconductivity type as that of the substrate, without using the N-typeelement formation region. The diode thus formed has a minimal area,eliminates parasitic element operation, and can also feed VSS to theP-type substrate.

Second, the diode connected to VDDQ is desirably formed on the N-typedevice element region by using the N-type well. The diode thus formedhas a minimal area, eliminates parasitic element operation, and can alsofeed VDDQ to the N-type element formation region.

Third, a diode other than the above two kinds of diode is desirablyformed directly on the P-type substrate by using the N-type well withoutforming any N-type element formation region. The diode thus formed has aminimal area and eliminates parasitic element operation.

FIGS. 25( a), 25(b), and 25(c) show further examples of the inter-powersupply protective device of this embodiment.

FIG. 25( a) shows an example of a particularly efficient way of forminga MOS transistor when the MOS transistor connections as shown in FIG.25( b) are made in a chip having a P-type silicon substrate and suppliedwith VSS. FIG. 25( c) shows a modification of the circuit of FIG. 25(a).

FIG. 25( a) shows a silicon substrate (P-type) 2401, an elementformation region (N-type) 2402, an N-type well 2403, a P-type well 2404,an N-type diffusion layer 2405, a P-type diffusion layer 2406, a gate2411, an N-channel MOS transistor 2407 on a P-type well formed on theP-type substrate, a P-channel MOS transistor 2408 on the N-type well2403 formed on the N-type element formation region 2402 (biased byVDDQ), an N-channel MOS transistor 2409 on the P-type well 2404 formedon the N-type element formation region 2402 (biased by VDDQ), and aP-channel MOS transistor 2410 on an N-type well formed on the P-typesubstrate 2401.

In the case of a chip where the silicon substrate is of P-type andsupplied with VSS, first, the N-channel MOS transistor connected to VSS,because it has the well of the same P-type as the substrate, isdesirably formed directly on the P-type substrate without forming anyN-type element formation region. The N-channel MOS transistor thusformed has a minimal area, eliminates parasitic element operation, andcan also feed VSS to the P-type substrate.

Second, the N-channel MOS transistor connected to VSSQ, though it has aP-type well, is desirably formed on the N-type element formation regionbiased by VDDQ. Thus, VSSQ can be fed to the P-type well of thisN-channel MOS transistor and be electrically isolated from the P-typesubstrate supplied with VSS, thereby eliminating parasitic elementoperation.

Third, an N-channel MOS transistor other than the above two kinds ofN-channel MOS transistors, although they have a P-type well, is formedon the N-type element formation region biased by VDD or VDDQ. Thus, VSSQcan be fed to the P-type well of this N-channel MOS transistor andelectrically isolated from the P-type substrate supplied with VSS,eliminating parasitic element operation.

Various modifications of the invention as set forth in the foregoingdescription will become apparent to those of ordinary skill in the art.All such modifications that basically rely on the teachings throughwhich the invention has advanced the state of the art are properlyconsidered within the spirit and scope of the invention.

1. A semiconductor integrated circuit device comprising: a plurality ofinput/output circuits arranged along by a chip end side of a chip onwhich the input/output circuits are formed; wherein each input/outputcircuit includes: a logic circuit; a level conversion circuit coupled tothe logic circuit; a pre-buffer coupled to the level conversion circuit;an NMOS output buffer and a PMOS output buffer coupled to an I/O pad andarranged to be driven by the pre-buffer; a first electrostatic breakdownprotective circuit coupled between the pre-buffer and the PMOS outputbuffer; a second electrostatic breakdown protective circuit coupledbetween the pre-buffer and the NMOS output buffer; a third electrostaticbreakdown protective circuit coupled between the logic circuit and thelevel conversion circuit; the PMOS output buffer is placed than thefirst electrostatic breakdown protective circuit near the I/O pad; andthe NMOS output buffer is placed than the second electrostatic breakdownprotective circuit near the I/O pad.